Reduction of aggregate EMI emissions of multiple transmitters

ABSTRACT

A power efficient and reduced electromagnetic interference (EMI) emissions multi-transmitter system for unshielded twisted pair (UTP) data communication applications. For each transmitter, digital transmit data is converted to a current-mode differential signal analog waveform by a digital-to-analog converter (DAC). The output current from each DAC is used to generate the required transmit voltage on the respective UTP line. Timing circuitry staggers the time base of each transmitter to reduce the aggregate EMI emissions of the multi-transmitter system.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application is a continuation of U.S. patent applicationSer. No. 09/429,988, filed Oct. 29, 1999, which claims the benefit ofthe filing date of U.S. Provisional Patent Applications Ser. Nos.60/106,265, filed Oct. 30, 1998 and entitled “POWER EFFICIENT ANDREDUCED EMI EMISSIONS TRANSMITTER”, 60/107,105, filed Nov. 4, 1998 andentitled “GIGABIT ETHERNET TRANSMITTER”, 60/107,702, filed Nov. 9, 1998and entitled “ETHERNET GIGABIT ANALOG SYSTEM”, and 60/108,001, filedNov. 11, 1998 and entitled “ADAPTIVE ELECTRONIC HYBRID LINE DRIVER FORGIGABIT ETHERNET”, the entire contents of which are hereby expresslyincorporated by reference.

BACKGROUND OF THE INVENTION

The present invention relates to transmission systems for transmittinganalog data on an unshielded twisted pair (UTP) of wires. Morespecifically, this invention is directed to an integrated gigabitEthernet transmitter.

The past few years has witnessed an almost exponential growth in theextent of high speed data networks, and the data transmission speedscontemplated over such networks. In particular, bidirectional datatransmission in accordance with the various Ethernet network protocols,over unshielded twisted pair (UTP) wiring, has emerged as the networkimplementation of choice for general commercial LAN installations aswell as for some of the more prosaic residential and academicapplications.

Local Area Networks (LAN) provide network connectivity for personalcomputers, workstations and servers. Ethernet, in its original 10BASE-Tform, remains the dominant network technology for LANs. However, amongthe high speed LAN technologies available today, Fast Ethernet, or100BASE-T, has become the leading choice. Fast Ethernet technologyprovides a smooth, non-disruptive evolution from the 10 megabits persecond (Mbps) performance of the 10BASE-T to the 100 Mbps performance ofthe 100BASE-T. The growing use of 100BASE-T connections to servers anddesktops is creating a definite need for an even higher speed networktechnology at the backbone and server level.

The most appropriate solution to this need, now in development, isGigabit Ethernet. Gigabit Ethernet will provide 1 gigabit per second(Gbps) bandwidth with the simplicity of Ethernet at lower cost thanother technologies of comparable speed, and will offer a smooth upgradepath for current Ethernet installations. With increased speed of GigabitEthernet data transmission, it is evident that EMI emission and linereflections will cause the transmitted signal to become substantiallyimpaired in the absence of some methodology for filtering thetransmitted data.

Therefore, there is a need for an integrated transmitter in a datatransmission system for pulse shaping digital input data and reducingEMI emissions, implemented with relatively simple circuitry.

SUMMARY OF THE INVENTION

A multi-transmitter communication system is configured for transmittinganalog signals over a multi-channel communication network. The system isconstructed to incorporate M transmitters, each having an output forserving a transmit signal on a transmit signal path electrically coupledbetween each communication channel and the output of the respectivetransmitter. A timing circuit is electrically coupled to eachtransmitter for providing the required timing signals for eachtransmitter. The timing signals for the transmitters define a clockdomain that is staggered in time resulting in a respective phase shiftof the output signals of each transmitter.

In one embodiment of the present invention, the timing signals arestaggered in time for predetermined time intervals to reduce aggregateelectromagnetic emission caused by signal images centered around integermultiples of frequency Fi of the M transmitters. M timing referencesstaggered in time by 1/(Fi*M) are generated by the timing circuit todrive the output of each of the M transmitters respectively. In anotherembodiment, each transmitter has a current-mode differential pair outputfor transmitting data on a UTP transmission line and the staggeredtiming is generated by a phase-lock-loop (PLL).

BRIEF DESCRIPTION OF THE DRAWINGS

The objects, advantages and features of this invention will become moreapparent from a consideration of the following detailed description andthe drawings in which:

FIG. 1 is a semi-schematic simplified block diagram representation of alocal and remote multi-transceiver system, in accordance with thepresent invention;

FIG. 2 is a semi-schematic, simplified block diagram of a transceiver,adapted for bi-directional communication, in accordance with the presentinvention;

FIG. 3 is a semi-schematic, simplified block diagram of the configurabletransmit DAC of FIG. 2;

FIG. 4 is a simplified functional diagram of a ROM including anintegrated digital filter and a DAC decoder;

FIG. 5 is a simplified block diagram of a multiple ROM embodiment;

FIG. 6 is a semi-schematic simplified block diagram of a multiple ROMembodiment;

FIG. 7 is a simplified block diagram of a ROM decoder;

FIG. 8 is a simplified block diagram of a ROM arrangement;

FIG. 9 is a semi-schematic simplified block diagram of a ROM decoder andrespective timing;

FIG. 10 is a simplified timing diagram for an integrated transmitter;

FIG. 11 is a simplified block diagram of one embodiment of aphase-locked loop;

FIG. 12A is a semi-schematic block diagram of switch logic circuitry forcontrolling operation of a DAC line driver current cell array;

FIG. 12B is a semi-schematic simplified block diagram of switch logiccircuitry and a line driver cell for a single current component;

FIG. 13 is a simplified schematic diagram of a DAC line driver cell,configured to operate in accordance with the present invention;

FIG. 14A is simplified schematic representation of Class-A switch logiccircuitry;

FIG. 14B is an exemplary truth table illustrating the operation of theClass-A switch circuitry of FIG. 14A;

FIG. 15A is simplified schematic representation of Class-B switch logiccircuitry;

FIG. 15B is an exemplary truth table illustrating the operation of theClass-B switch circuitry of FIG. 15A;

FIG. 16 is a simplified block diagram of an analog discrete-time filterand a line driver cell;

FIG. 17 is a schematic representation of one implementation of a delaycell;

FIG. 18 is a simplified timing diagram of a signal before and afterdiscrete-time filtering;

FIG. 19 is a semi-schematic block diagram of one implementation of ananalog output filter;

FIG. 20 is a schematic representation of one implementation of an analogoutput filter;

FIG. 21A is a simplified timing diagram of a signal before discrete-timefiltering;

FIG. 21B is a simplified timing diagram of the signal in FIG. 21A afterdiscrete-time filtering;

FIG. 22 is a semi-schematic, simplified block diagram of one arrangementof an integrated transceiver including transmission signal cancellationcircuitry and a simplified line interface, in accordance with thepresent invention;

FIG. 23 is a semi-schematic, simplified circuit diagram of oneimplementation of a precision bias current generator for the transmitDAC of FIG. 22;

FIG. 24 is a semi-schematic, simplified circuit diagram of oneimplementation of a variable bias current generator for the replica DACsof FIG. 22;

FIG. 25 is a simplified timing diagram illustrating transmission signalperturbation of a receive signal and the effects of transmission signalcancellation in accordance with the present invention;

FIG. 26 is a simplified block diagram of multiple transmittersconfigured for reduction of aggregate emissions, in accordance with thepresent invention; and

FIG. 27 is simplified timing diagram of the image component of afour-transmitter system.

DESCRIPTION

In many transmission system, the signal to be transmitted over atransmission line is processed and filtered to minimize signaldistortion and Electromagnetic Interference (EMI) emission in thetransmission line. Typically, this wave-shaping and filtering is carriedout digitally for more accuracy. Therefore, the digital signal need tobe converted to an analog signal, for transmission over the UTPtransmission line, using a Digital-to-Analog Converter (DAC) .Conventionally, digital signal processing and digital filtering iscarried out separately and then, the “shaped” digital signal isconverted to analog signal.

Generally, a DAC includes an array of output driver cells controlled bya DAC decoder. The DAC decoder generates control words responsive to thedigital input. The control word controls each output driver cell byturning the current of a respective output driver ON or OFF. An analogsignal is generated by connecting all of the outputs of the drivercells. This method generally requires additional circuits and speciallogic circuits for implementing the DAC decoder and re-synchronizationlogic to re-synchronize the bits in a control word for driving all ofthe output driver cells at the same time. The requirement for theseadditional circuits becomes even more significant and problematic in anIntegrated Chip (IC) where silicon area is expensive. It would bebeneficial, both to circuit performance and to manufacturing economies,if the digital filter and the DAC decoder in a data transmission systemcan be integrated in a memory device such as a Read-Only Memory (ROM).

Furthermore, a conflict arises when it is recognized that radiativeemissions are reduced when a differential signal transmitter, such as anEthernet transmitter, is transmitting a differential signal in what istermed Class-A mode, i.e., the differential mode current varies in orderto define the signal, while the common-mode current component is keptconstant. However, constant common-mode current compels such circuitryto conduct a constant quanta of current at all times, even when thedifferential mode signal defines a zero value. It is well understoodthat current mode transmitters, outputting a constant common-modecurrent, necessarily consume relatively large amounts of power, causedby constant conduction of the output section. It is further understoodthat in order to minimize constant current conduction and thus powerconsumption, a differential signal system could be operated in what istermed a Class-B mode, i.e., one in which the common-mode current isallowed to vary between some maximum value and zero. However, whenoperating in Class-B mode, the variable common-mode current causes thevery radiative emissions that one would seek to avoid in a high densityinstallation.

It is beneficial, therefore, both to circuit performance and tomanufacturing economies, if an Ethernet-capable transceiver includes atransmitter or transmit DAC that was adaptively configurable to operateas a cross-standard transmitter platform, as well as being adaptivelyconfigurable between Class-A and Class-B operational modes, depending onthe intended installation. Such a circuit provides the industry with asingle-chip solution having such flexibility that it is able to beincorporated into high density systems where emissions are a problem, aswell as low density systems where power consumption is the greatestconcern. Such a single-chip solution is able to communicate with otherEthernet installations regardless of the communication standard chosen.

As the number of available communication channels increases, moretransmitters need to be integrated in an IC chip or in a Printed CircuitBoard (PCB). With increasing speed of circuits and clock rates, it isevident that EMI emission will cause the transmitted signal to becomesubstantially impaired in the absence of some methodology to reduce theemission.

The output spectrum of a differential current-mode transmission linedriver includes signal harmonics radiating from commonly employedtransmission media such as UTP cable. A transmission line driver, evenwith filtering, includes these signal harmonics having substantial powerdensity. The harmonics have images of the baseband signal centeredaround the integer multiple frequencies of the interpolation rate N. Forexample, for an input data rate of 1/T, the harmonics are centeredaround 1*N/T, 2*N/T, 3*N/T, . . . . The differential energy producedfrom these images is converted to common-mode energy by the finitedifferential-to-common-mode conversion in the magnetic and UTP medium.The transmitted common-mode energy is the primary source of EMIemissions for data communication applications. These EMI emissions maygenerate crosstalk between system components or cause errors in datatransmission.

The first set of images around N/T is the highest of the images and isthe major contributor to EMI emissions. For example, images of thebaseband signal in 10Base-T transmission medium with a 20 MHztransmission rate and interpolation rate of 8 are centered around 160MHz, 320 MHz, 480 MHz, . . . . The highest image is centered around 160MHz and significant baseband energy is located at 150 MHz and 170 MHz(i.e. 160 MHz +/−10 MHz).

This EMI emission becomes even more significant and problematic in datatransmission systems such as IC chips that integrate severaltransmitters in a single chip. In these applications, a furtherfiltering of the output waveform is required in order to meet theFederal Communications Commission (FCC) emission requirements that limitthe magnitude of signal harmonics which may be radiated by a givenproduct.

It is known in the art that EMI emissions induced by a transmitter in adata transmission system can be reduced a by cancellation circuit forgenerating a cancellation signal to produce electromagnetic fields whichare opposites of the fields produced by the transmitters. This methodgenerally requires additional circuits for adjusting the phase andamplitude of the cancellation signal. Thus, the method is costly andcumbersome, specially, for data transmission systems that includemultiple transmitters.

It would be beneficial, both to circuit performance and to manufacturingeconomies, if the EMI emission in a multi-transmitter system is reduced,without the need for complex and costly cancellation circuitry. Such EMIreduction can be accommodated by circuitry resident on amulti-transmitter chip or on a multi-transmitter PCB.

Moreover, it is known in the art that emission induced by a transmissionline can be reduced by wave shaping employing digital filtering methods.The effectiveness and pulse shaping quality of a digital filter dependon its interpolation rate. However, the higher the interpolation rate,the more complex the digital filter gets. Thus, utilizing a combinationof a simpler digital filter with a lower interpolation rate and ananalog discrete-time filter, instead of a more complex digital filterwith twice the interpolation rate of the simpler digital filter,achieves similar performance resulting in a significant reduction indigital filter complexity and size. In an IC implementation, thereduction of the interpolation rate of the digital filter, results insignificant decrease in silicon area and power consumption of thetransmitter.

Additionally, the latest high-speed Ethernet protocols contemplatesimultaneous, full bandwidth transmission, in both directions (termedfull duplex), within a particular frequency band, when it is desirableto maximize transmission speed. However, when configured to transmit infull duplex mode, it is evident that the transmitter and receiversections of a transceiver circuit must be coupled together , in parallelfashion, at some transmission nexus short of twisted pair transmissionchannel.

Because of the nexus coupling together of the transmitter and receiver,it is further evident that the simultaneous assertion of a receivesignal and a transmit signal, on the transmission nexus, will cause thereceive signal to become substantially impaired or modified in theabsence of some methodology to separate them.

Standard arrangements for achieving this isolation or transmit/receivesignal separation in the prior art include complex hybrid circuitryprovided as a separate element external to an integrated circuittransceiver chip. Hybrids are generally coupled between thetransmit/receive signal nexus (the channel) and the transmit and receivesignal I/Os. In addition to excess complexity and non-linear response,hybrid circuits represent costly, marginally acceptable solutions to thetransmit/receive signal separation issue.

It would be beneficial, both to circuit performance and to manufacturingeconomies, if a local transmit signal is separated from a receivesignal, in full duplex operation, without the need for complex andcostly hybrid circuitry. Such separation is accommodated by circuitryresident on an integrated circuit transceiver chip and in relativeproximity to the signals being processed. Such separation is furtherperformed in a substantially linear fashion, i.e., frequencyindependent, and be substantially immune to semiconductor processtolerance, power supply and thermal parameter variations.

The present invention might be aptly described as a system and methodfor an integrated data transmission system for pulse shaping digitalinput data, generating synchronized DAC control signals, and reducingEMI emissions in such a way to simplify the complexity of circuits andincrease the flexibility of the system. The invention contemplates amemory device, such as a ROM, including data implementing the functionsof a digital filter and the functions of a DAC decoder combined. DACline driver cells are adaptively configurable to operate in either aclass-A or a class-B mode depending on the desired operational modality.A discrete-time analog filter is integrated with the DAC line driver toprovide additional EMI emissions suppression. An adaptive electronictransmission signal cancellation circuit separates transmit data fromreceive data in a bidirectional communication system operating in fullduplex mode. For a multi-transmitter system, timing circuitry staggersthe time base of each transmitter to reduce the aggregate EMI emissionsof the multi-transmitter system.

FIG. 1 is a simplified block diagram of a multi-pair communicationsystem that includes an integrated digital filter and DAC decoder (notshown), an adaptively configurable Class-A/Class-B circuitry 10, adiscrete-time analog filter 9, an adaptive transmission signalcancellation circuitry 5, and a staggered timing generator 7 for EMIreduction, according to one embodiment of the present invention. Thecommunication system illustrated in FIG. 1 is represented as apoint-to-point system, in order to simplify the explanation, andincludes two main transceiver blocks 2 and 3, coupled together with fourtwisted-pair cables. Each of the wire pairs is coupled betweenrespective transceiver blocks and each communicates informationdeveloped by respective ones of four transmitter/receiver circuits(constituent transceivers) 6 communicating with a Physical CodingSublayer (PCS) block 8.

Each transmitter circuit is coupled to a respective wire pairtransmission media. Although FIG. 1 illustrates a single driver circuitcorresponding to a respective twisted wire pair, the illustration issimplified for ease of explanation of the principles of the invention.It should be understood that the transmitter within each transceiver 6represents a multiplicity of differential output cells, the sum of whichdefines the physical signals directed to the transmission medium.

The functions of a digital filter, a DAC decoder, and are-synchronization logic are combined in a memory device, such as a ROM.The timing generator circuit 7 provides timing references for amultiplexer and the respective control logic for time multiplexing theoutput of the memory device. This allows a transmitter system,constructed according to the present invention, to operate mostefficiently in a reduced circuit complexity and silicon area.

Adaptively configurable Class-A/Class-B circuitry 10 allows forselective low-power and/or high-speed operation. A selection circuitasserts control signals that adaptively configure each signal componentoutput circuit to operate in Class-A, Class-B, or a combination ofClass-A and Class-B mode.

An analog discrete-time filter 9 is implemented for reducing EMIemission at the output of the transmitter. In one embodiment, timinggenerator circuit 7 generates timing signals for dividing each digitizedinput data sample into a first time segment and a second time segment. Acontrol logic connected to the output cell generates control signals todrive the output cell to produce half of the current-mode differentialoutput signal for the first time segment and the full current-modedifferential output signal for the second time segment.

A transmit signal cancellation circuit 5 is electrically coupled to thereceive signal path, and develops a cancellation signal, which is ananalogue of the transmit signal, and is asserted to the receive signalpath so as to prevent the transmit signal from being superposed on areceive signal at the input of the receiver.

The timing signals for each transmitter are staggered in time forpredetermined time intervals to reduce aggregate electromagneticemission caused by signal images centered around integer multiples offrequency Fi of the four transmitters. Each transmitter circuit iscoupled to a timing generator circuit 7 which provides the requiredtiming for the respective transmitter in accordance with the presentinvention.

FIG. 2 is a simplified block diagram of one implementation of atransceiver system, adapted for full-duplex communication, thearrangement of which might be pertinent to an understanding of theprinciples of operation of the present invention. The exemplarytransceiver of FIG. 2 encompasses the physical layer (PHY) portion of atransceiver and is illustrated as including a transmitter section 30 anda receiver section 32, coupled between a media access layer (MAC) 20 anda communication channel; in this case, represented by twisted pairwiring 4, also termed unshielded twisted pair (or UTP) wiring.

The transceiver of the illustrated embodiment operates in accordancewith a transmission scheme which conforms to the 1000BASE-T standard for1 gigabit per second (Gb/s) Ethernet full-duplex communication over fourtwisted pairs of Category-5 copper cables. For ease of illustration anddescription, the embodiment of FIG. 2 depicts only one of the four 250Mb/s constituent transceivers which are configured in parallel fashionand which operate simultaneously to effect 1 Gb/s in order to effect 1Gb/s communication. Where signal lines are common to all four of theconstituent transceivers, they are rendered in a bold line style. Wheresignal lines were laid to a single transceiver, they are rendered in athinner line style.

Received analog signals are provided to the receiver section 32 wherethey may be pre-conditioned by filter/amplification circuitry 457, suchas a high-pass filter (HPF) and programmable gain amplifier (PGA),before being converted to digital signals by a receive analog-to-digitalconverter (ADC) 56 operating, for example, at a sampling rate of about125 MHz. ADC timing is controlled by the output of a timing recoverycircuit 58 which might be configured as a phase-lock-loop (PLL) or someother feed-back controlled circuitry configured for determinableperiodic operation.

Digital signals, output by the receive ADC 56, along with the outputsfrom the receive ADCs (not shown) of the other three constituenttransceivers, are input to a pair-swap multiplexer circuit (MUX) 55which functions to sort out the four input signals from the four ADCsand direct each signal to its respective appropriate demodulator circuitfor demodulation and equalization. Since the coding scheme for gigabitcommunication is based on the premise that signals carried by eachtwisted pair of wire correspond to a 1-dimensional (1D) constellationand that the four twisted wire pairs collectively form a 4-dimensional(4D) constellation, each of the four twisted wire pairs must be uniquelyidentified to a particular one of the four dimensions in order thatdecoding proceed accurately. Any undetected and uncompensated swappingof wire pairs would result in erroneous decoding. The pair swap MUX 55maps the correct input signal to the demodulation circuit 28.

Demodulator 28 functions to demodulate the receive digital signal andmight also provide for channel equalization. Channel equalization mightsuitably include circuitry for compensating theinter-symbol-interference (ISI) induced by partial response pulseshaping circuitry in the transmitter section of a remote gigabit capabletransceiver, which transmitted the analog equivalent of the digitalreceive signal. In addition to ISI compensation, the demodulation alsocompensates for other forms of interference components such as echo,offset and near end cross-talk (NEXT) by subtracting correspondingcancellation vectors from the digital receive signal. In particular, anoffset cancellation circuit 27 generates an estimate of the offsetintroduced at the transceiver's analog front end (including the PGA andADC).

Three NEXT cancellation circuits, collectively identified as 26, modelthe near end cross-talk impairments in the receive signal caused byinterference between the receive signal and the symbols (signals) sentby the transmitter sections of the other three local constituenttransceivers. Since the NEXT cancellation circuits 26 are coupled to thetransmit signal path, each receiver has access to the data transmittedby the other three local transmitters. Thus, NEXT impairments may bereplicated by suitable filtering. By subtracting the output of the NEXTcancellation circuits 26 from the receive signal, NEXT impairments maybe approximately canceled.

Following echo, NEXT and offset cancellation, receive signals aredecoded (by a trellis decoder, for example) and provided to a receivePhysical Coding Sublayer (PCS) lock 24 and thence to the media accesslayer (MAC) 20 through a media independent interface circuit (GMII) 23.

In transmit operations, transmit signals are provided by the MAC 20 to atransmit PCS block 22 through a transmit GMII circuit 21. In the case ofgigabit Ethernet transmissions, coded signals might be processed by apartial response pulse shaping circuit (not shown) before being directedto a transmit digital-to-analog converter (TXDAC) 29 for conversion intoanalog signals suitable for transmission over twisted pair wiring 4 to aremote receiving device through line interface circuitry 59.

The exemplary transceiver system of FIG. 2 has been described in thecontext of a multi-pair communication system operating in conformancewith the IEEE 802.3 standard (also termed 1000BASE-T) for 1 gigabitEthernet full-duplex communication over Category-5 twisted pair wiring.However, and in accordance with the present invention, the exemplarytransceiver is further configurable for operation in conjunction with10BASE-T, 100BASE-T and 100BASE-Tx performance standards. In particular,the transmitter 29 is configurable to accommodate both 1.0 volt outputswings characteristic of Tx and the 2.5 volt output swingscharacteristic of 10BASE-T operation.

Bidirectional analog signals are transmitted to and received from a2-wire transmission channel 4 through line interface circuitry 59. Inthe illustrated transceiver system of FIG. 2, both the transmitter 30and receiver 32 are coupled to the transmission channel 4 through theline interface circuitry 59 such that there is a bidirectional signalpath between the transceiver and the interface circuit 59. Thisbidirectional signal path splits into a receive signal path and atransmit signal path at a nexus point 64, at which point both transmitand receive signals are present during full duplex operation. Transmitsignals, present on the nexus 64, are isolated from the receive ADC 56by a transmit signal cancellation circuit 5 which is coupled between thebidirectional signal nexus and the receiver's analog front end.

In a manner to be described in greater detail below, transmit signalcancellation circuitry 5 functions to evaluate signals appearing on thereceive signal line and condition those signals such that any transmitsignal components are removed from the receive signal line prior to thereceive signal's introduction to the analog front end and the receiveADC 56. Further, such conditioning does not perturb any components ofthe transmit signal prior to the signal's introduction to the channel.Transmit signal cancellation circuitry 5 is connected to receive, and isoperatively responsive to, the digital transmission signal directed tothe transmit DAC 29 by the pulse shaper 22. Since the cancellationcircuit 5 operates in response to the same digital transmission signalas a transmit DAC 29, the cancellation circuit 5 is able to develop aconditioning or cancellation signal which substantially directlycorresponds to the analog transmission signal produced by atransceiver's transmit DAC.

In general terms, any analog intelligence signal, whether in baseband orpassband, may be processed by the cancellation circuit 5 for full duplexcommunication over any transmission channel. However, the intelligencesignal characteristics are effectively canceled at the inputs of thereceive ADC 56 such that full duplex communication can occur without atransmitter's intelligence signal swamping a receive signal that mighthave been communicated over a generally lossy channel, characterized bya relatively poor noise margin or signal-to-noise ratio (SNR) . Thetransmit intelligence signal is conditioned prior to its being directedto the transmission channel, thus allowing the system to operate on acleaner signal, resulting in a cleaner, more effective and precisesignal suppression characteristic at the receive end of the nexus.

In other words, the cancellation circuit 5 is positioned at a nexusjunction of a bi-directional transceiver's transmit block, receive blockand transmission channel buffer circuitry, as represented by a lineinterface circuit. The cancellation circuit operates upon transmitsignals appearing on the nexus so as to allow substantially unperturbedpassage of analog transmit signals to the channel side of the nexus,while restricting passage of analog transmit signals to the receive sideof the nexus such that receive signals can be processed by the analogfront end unimpaired by superposed components of transmit signals.

Timing circuit 7 generates the required timing for the plurality oftransmitters. In a manner to be described in greater detail below, eachtransmitter 29 is constructed to include a digital-to-analog converter(DAC) with an array of output driver cells, with individual cells makingup the array able to be adaptively included or excluded from operationin order to define a variety of characteristic output voltage swings.The individual output driver cells are controlled by a DAC decoder.Responsive to the value of the digital input, the DAC decoder generatesa DAC control word that controls which sets of output cells are turnedon and which sets are turned off.

The output current of the DAC is generated by an array of identical linedriver cells, each with respective driver controls coming from a DACdecoder. For each value of the digital input, the DAC decoder generatesa control word. Depending on the DAC control words, these driver cellsare either turned on or turned off. For each digitized sample of theinput, the output currents of all the line driver cells are addedtogether to produce an analog representation of the digital input. Thenumber of line driver cells is chosen to meet the resolution requirementof the DAC. Each line driver cell has high output impedance, such thatthe transmit output impedance of the transmitter is determined by anexternal resistor. All driver cells have topologically identical circuitdesign, so each transmitter line driver can achieve accurate and linearoutput current levels.

FIG. 3 shows one embodiment of a transmitter 29 architecture. Thetransmitter includes an interpolating digital filtering function forpulse shaping of the transmit signal to reduce the EMI emission causedby the transmission line. Pulse shaping includes modification of asignal spectrum by reducing the sharp edges of the signal and iseffective in lowering EMI emissions within a transmission system. A DAC(not shown as a separate block) converts the filtered digital output toan analog signal current.

Input digital data is fed to an interpolating digital filter 33. Thefiltered data then goes to a DAC binary decoder 34, which produces theDAC control words. Each bit in a control word controls an output drivercell by turning the current cell ON or OFF. The control words aredirected to DAC current-mode line driver array 36 which includes anumber of output driver cells. The outputs of all the current cells areadded together to create the output analog signal. The number of drivercells is determined by the desired resolution of the DAC. Theinterpolating function of the digital filter 33 is integrated with thebinary decoding function in a memory device, such as ROM 31. In otherwords, the functions of the digital filter and the DAC decoder areimplemented as part of the ROM content. This ROM replaces digitalfiltering circuits, DAC decoding logic, and re-synchronization logic.When implemented in such manner, the logical implementation and memoryreplaces digital filtering circuits, DAC decoding logic circuit andre-synchronization logic circuits that are conventionally implemented inhardware. Thus, the hardware functionality of these circuits is renderedinto arithmetic form and implemented in a memory device.

The output data of the ROM (filtered and decoded data) is selected by amultiplexer 35 that is synchronized employing a time reference 7.Re-synchronization logic that is usually needed at the output of a DACdecoder and is generally integrated with a DAC line driver in the art ofDAC design is no longer needed because the DAC decoding function isperformed in the ROM and is subsequently synchronized by the multiplexer35. A stable and well-controlled timing reference 7 generates thecontrol clocks and timing delays to the various blocks from a masterclock.

The output of the multiplexer is further filtered by a discrete-timeanalog filter 9. The discrete-time analog filter is integrated with theDAC line driver array 36 to suppress high-frequency harmonics of theoutput transmit signal. Depending on the output of the multiplexer, aselected number of current drivers in the line driver array 36 areturned on to produce a current corresponding to the value of thefiltered digital input signal. The line driver array produces adifferential current output that drives the UTP line load. The linedriver array 36 can be controlled for a power efficient operation usingthe adaptively configurable class-A/class-B circuit. In one embodimentof the present invention, an analog output filter 37 further processesthe output signal from the line driver for smoother edges to furtherreduce the EMI emissions.

In one embodiment, the digital filter 33 is a Finite Impulse Response(FIR) filter. The output of a FIR filter is a weighted sum of thepresent and past input samples only, and is not a function of theoutput. To perform an interpolation function for wave shaping of thetransmit signal, a weighted sum of the present and past input signals iscalculated to produce the output of the filter. The weighted sum isdetermined by selection of filter coefficients. The order of theprevious inputs that are taken into account for determining a presentoutput is called the order of the filter.

FIG. 4 shows a functional diagram of the ROM 31 including the digitalfilter 33 and DAC decoder 34. The digital filter function is partitionedinto N smaller digital filters 46 a-46 h which operate at the input datarate 1/T but are staggered by 1/Nth of the data period. In other words,with an interpolation rate N of eight, there are eight smaller digitalfilters. Each smaller filter is essentially a smaller ROM. Conceptually,the input data goes to two shift registers 41, 42 for an exemplarysecond order filter. For each smaller filter, the respective previousinput data strings are multiplied by the respective filter coefficientsC0-C15 and then added to generate the output for each smaller filter.The outputs of the smaller filters are fed to a respective DAC decoder43 a-43 h. For example, in filter #0, the data strings are multiplied bythe coefficients C0 and C8 and added before going to the DAC decoder 43a. Inside the ROM, the shift registers and the digital filters becomeselection circuits for selecting the respective ROM word. In oneembodiment, the interpolation of the digital signal is performed by afunctional twenty four order filter, implemented by eight functionalthird-order filters in a ROM including three shift registers.

Referring back to FIG. 4, the eight outputs of the digital filters 26a-26 h are processed by eight binary decoders 43 a-43 h, which convertthe outputs to DAC control words 47 a-47 h. The 8-to-1 multiplexer 35selects one of the DAC control words at 8 times the data rate so, themultiplexer output rate is 8/T. For the example in FIG. 4, in 10Base-T,N is 8 and the DAC control word rate is 8 times 20 MHz or 160 MHz. Thetiming between the multiplexer selection control 45 and digital filteroperation allows sufficient settling time for each filter and decodercombination.

For other interpolation rates N, there are N digital filters and Nbinary decoders to produce N control words. An N-to-1 multiplexerselects control words at N times the data rate to provide a multiplexeroutput rate of N/T.

The selection control and ordering of the digital filters follows a Graycode ordering which prevents glitches in the DAC control word becausethe selection only allows transitions to the proper subsequent filter. AGray code is a binary code in which sequential numbers are representedby binary expressions, each of which differs from the precedingexpression in one place only. In addition, the Gray coded selectioncontrol has the feature that no control bit lines are required tooperate higher than half the multiplexer selection rate, i.e., 0.5*N/T.Since the DAC control word is synchronized by the multiplexer controlselection, a bank of re-synchronization latches is not needed in theDAC. The eight filters in FIG. 4 are pictorially arranged to illustratea Gray coding selection by the multiplexer 35.

The input data rate of the digital filter 33 is 1/T where T is, forexample, 40 ns for 100Base-T4 and 50 ns for 10Base-T Ethernetcommunication lines. The input data is interpolated by the rate N. Theinterpolating digital filter produces output samples at N/T. Thecoefficients of the filter are chosen to meet the pulse shaperequirement of the particular communication application. For example, in10Base-T, the coefficients follow a linear filter which produces a 100%raised cosine response after it has been filtered by a 100 meter UTPline model. In 100Base-T4, the coefficients follow a linear filter whichproduces a 100% raised cosine response after it has been filtered by athird order Butterworth filter.

The digital filter is designed to meet the input signal requirements ofa particular communication line. The coefficients of the filter arechosen by looking backwards to determine what values for the filtercoefficients would produce the desired output signal. For example, a100% raised cosine response is required in one embodiment for a 10Base-Ttransmission line and the filter coefficients are selected based on thetransfer function of the transmission line and the required output. Thefilter results are then saved in a ROM as look up tables. In otherwords, the coefficients are used to determine the content of the ROM.The DAC decoder function is integrated and saved in the same lookuptable in the ROM, along with the coefficients of the digitalinterpolating filter. As a result, every word of the ROM includes allthe functions for computing the filter output as well as all thefunctions for decoding the DAC. This technique not only eliminates theneed for a separate digital filter circuit, but also eliminates the needto re-synchronize the output of the DAC decoder before it goes to a DACdriver cell.

A phase-locked loop (PLL) is used to generate the required timingsignals (time reference 7) for outputting the right data at the righttime from the ROM. A transmitter that supports multiple communicationapplications such as a 10Base-T, 100Base-T4/TX/T2, or 1000Base-Tproduct, requires different digital filtering (e.g., different valuesfor the filter coefficients). Thus, multiple smaller ROMs (digitalfilters) are implemented, but only the output from the appropriatesmaller ROM is selected by using a transmission mode control signal.FIG. 5 shows an exemplary embodiment for 10Base-T, 100Base-TX, and1000Base-T communication modes. Depending on the transmission mode, amode select control selects one of the three smaller ROMs 51, 52, or 53,and the output of the selected ROM goes to the multiplexer. The twoother smaller ROMs that are not selected are inactive and thusdisconnected from the output line.

There are as many rows in each smaller ROM as there are bits in the ROMword. For example, a ROM word of j bits has j rows. Also, there are iwords stored in each smaller ROM. Referring now to FIG. 6, when the10Base-T mode is selected, ROM 51 is active and ROMs 52 and 953 areinactive and disconnected from the output line. Specifically, all theMOSFETs, Mbij and Mcij, are off and thus are floating. Depending on thecontent of the ROM 51, the 10Base-T control 61 may turn on one of theMOSFETs Mall-Mali in row 1, resulting in a low logic level at theoutput. The MOSFETs in other rows of the ROM would be open or closedaccordingly as required by the ROM word.

FIG. 7 illustrates one embodiment of the ROM control logic for athree-tap filter implementation. The input data is shifted by threeshift registers 71, 72, and 73 that are clocked by PHI1 running at 40MHz to produce the ROM control signals Q0, Q1, and Q2. However, threemore shift registers 72, 74, and 76 that are clocked by PHI1B (PHI1inverted) are used to generate three more ROM control signals Q0 d, Q1d, and Q2 d. These two sets of ROM control signals, one set delayed intime, are used to generate the two halves of a ROM word at two differenttimes. This technique ensures that there is sufficient time for settlingof the ROM data at the input of the multiplexer 35.

Each smaller ROM included in the ROM 31 can be organized as several ROMarrays, each ROM array having a different timing for outputting the ROMdata. As shown in FIG. 8, each smaller ROM is divided into two ROMarrays. The first ROM array contains data cells for the first half ofeach ROM word (O(0-3)) and is controlled by the ROM decoder 81 that usesQ0-Q2 control signals. The second ROM array contains data cells for thesecond half of each ROM word (O(4-7))and is controlled by the ROMdecoder 82 that uses Q0 d-Q2 d control signals. Thus, O(0-3) aresynchronized to PHI1 and O(4-7) are synchronized to PHI1B to ensuresufficient data settling time. The 8-to-1 multiplexer 35 selects each ofthe ROM word bits O(0-7) based on a Gray code ordering to ensure furtherintegrity of the signals going to the DAC decoder.

A block diagram of an exemplary ROM decoder and timing signals for a twobit input data for each transmitter is depicted in FIG. 9. Two clockphases CK0 and CK4 and their inversions CK0B and CK4B are generated froma PLL (shown in FIG. 11). These clock phases are buffered by an inputclock buffer 91 before they are fed to an FIR clock generator 92. Basedon clock phases MCK0 and its inversion MCK)B, and MCK4 and its inversionMCK4B, the clock signals PHI1 and PHI1B are generated by the clockgenerator 92. The clock signals PHI1 and PHI1B are used by the register93 to generate Q0-2 and Q0 d-2 d ROM control signals. These controlsignals are then fed to the ROM 31.

The timing diagram of the ROM 31 and the multiplexer 35, for aninterpolation rate of eight, is shown in FIG. 10. The clock signal PHI1is generated from clock phase MCK0 and is used to clock the input datato produce Q0-2 ROM control signals. PHI1B, the inversion of PHI1, isused to clock the input data to generate Q0 d- 2 d ROM control signals.The ROM control signals Q0-2 are used to generate ROM outputs O(0-3) andQ0 d-2 d are used to generate ROM outputs O(4-7). Multiplexer selectsignals SEL0-2, following a Gray coding scheme, are used to multiplexthe ROM outputs at eight times the frequency of the MCK0.

The timing signals can be accurately generated by timing generatorcircuit, such as a PLL that includes a Voltage Control Oscillator (VCO).When a PLL is used as a frequency synthesizer, the VCO is divided downto a reference frequency that is locked to a frequency derived from anaccurate source such as a crystal oscillator. FIG. 11 shows a PLL usedfor generating the required timing signals for one embodiment of thepresent invention. Phase detector 111 produces two periodic outputsignals as a function of the difference in the frequencies of its twoinput clocks. These two outputs are fed to a charge pump 112. The outputof the charge pump 112 has a tri-state capability. Depending on whichinput is turned on, the output of the charge pump is a positive currentsource, negative current source, or an open circuit.

A filter 113 filters the high frequency components of the output of thecharge pump before it is inputted to a VCO 114, in order to keep the VCOstable. The output of the VCO is divided by five (115) such that itlocks to the crystal oscillator before it 35. is fed back to the phasedetector 111 as its first input. The second input of the phase detectoris driven by a master clock. This way, clock signals at a multiple ofthe master clock are created. Selection and ordering of the DAC decoderoutput through the MUX follows a Gray-code selection criteria whichprevents glitches in developed DAC control words because the selectioncriteria only allows transitions to proper decoder outputs.

FIG. 12A is a semi-schematic block diagrammatic representation ofClass-A/B switch logic circuitry 120, suitable for receiving a DACcontrol word and generating a plurality of line driver cell controlsignals, each set of control signals corresponding to a particular oneof individual line driver cells making up a line driver array. DACcontrol words control operation of a Class-A/B switch logic circuit 120which, in turn, provides activation signals to individual line drivercells making up a line driver array 122. Characteristically, the outputcurrent of a DAC is generated by an array of identical line driver cellswhich are turned-on or turned-off depending on the state of a particularDAC control word. For each input sample, output currents of all of theactive line driver cells are added together at a summing junction toproduce an analog representation of the original digital input. Controlof individual driver cells and their operational mode (Class-A/B) isdetermined by “select” signals provided to the Class-A/B switch logiccircuit 120. Necessarily, the number of the individual line driver cellsimplemented and their characteristic operational mode is chosen in orderto meet the resolution requirements of the DAC as defined by thetransmission standard.

For a transmitter that supports multiple communication standards such as10BASE-T, 100BASE-T4/Tx/T2, 1000BASE-T, and the like, the number ofindividual driver cells making up the driver array will depend on themaximum, worst-case output voltage swing required by the transmissionstandards. In the exemplary embodiment, there are twenty-five individualcurrent driver cells, each outputting a particular current quanta andfor purposes of this specification, will be deemed normalized such thateach of the twenty-five cells might be termed “full” cells. In addition,the line driver array 122 includes a “half” cell, so defined because thecurrent quanta produced by that cell exhibits a value one-half the valueof the current quanta output by the twenty-five “full” cells.Accordingly, depending upon the actual value of the current quanta andthe load across which the output current is developed, full value outputswings can be developed by the transmitter of the present invention infifty equal-sized “half” steps by switching various combinations of“full” cells and the “half” cell into operation.

For example, in normal 10BASE-T operation, the output voltage swingdefined by the standard is 2.5 volts. In order to accommodate thisoutput voltage swing, all twenty-five cells, plus the “half” cell areused to develop the output. It will be understood by those having skillin the art that each of the twenty-five “full” cells develops a currentsufficient to develop 0.10 volts across a load, with the “half” cellproviding an additional degree of granularity to the output. Conversely,in 100BASE-Tx mode, the standard defines a 1.0 volt output swing. Withdriver cells configured to each develop 0.10 volts across a load, onlyten cells are required from the line driver array in order toaccommodate this output swing.

In FIG. 12A,the switch logic circuit 120 includes twenty-six Class-A/Bcontrol circuits 122 each of which defines whether their respective linedriver cell is operable or non-operable and, if operable, whether eachcorresponding driver cell outputs a differential current in Class-A orClass-B mode. Each of the Class-A/B control circuits 122 defines fouroutput signals a, b, c and d which, in a manner to be described furtherbelow, controls both operation and mode of each line driver cell.Control signals are asserted by each of the control circuits 122 inaccordance with a select signal (SEL) asserted by the timing reference 7of FIG. 3.

Turning now to FIG. 12B, in one embodiment of the present invention,each current drive cell 126 is able to be controlled for either Class-A,Class-B, or a combination of Class-A and Class-B operation by selectingcontrol signals a, b, c and d from either a Class-A driver control logiccircuit 123 or a Class-B driver control logic circuit 124 by a 2:1 MUX125. Determination of whether the line driver cell will be driven inClass-A or Class-B mode is made by a select signal that determines whichof the control signals (a, b, c and d) will be selected by the MUX 125.Further, determination of the binary state of the control signals (a, b,c and d) is made by two input signals In0 and In1 which make up thatportion of the DAC control word directed to that particularcorresponding Class-A/B switch logic section. An exemplary adaptivelyconfigurable Class-A/Class-B circuit is described in detail below.

It should be noted here that the DAC decoder 34 (FIG. 3) willnecessarily have as many outputs as there are individual line drivercells to be driven, i.e., the output of the DAC decoder is 26 wide inthe exemplary embodiment. Thus, the DAC decoder is capable of providingtwenty-six pairs of In0 and In1 control signals; one pair directed toeach switch logic and line driver cell combination.

Turning now to FIG. 13, an exemplary embodiment of an individual linedriver cell is indicated generally at 126. In general terms, the linedriver cell 126 might be aptly described as two differential pairscross-coupled to define a differential output (IpIn). Current flowingthrough each of the differential pairs is defined by two n-channelcurrent source transistors 131 and 132 each of which have their gateterminals coupled to a stable bias voltage developed by an n-channeltransistor 133 configured as a voltage follower. The bias voltagegenerated by the MOSFET diode transistor 133 is determined by thecharacteristic value of a current source 138 which provides a stablecurrent reference to the MOSFET diode transistor 133 such that a stablebias voltage is developed on its gate terminal.

As is well understood in the art, the current source transistors 131 and132 conduct a characteristic current which is proportional to thecurrent developed by the current source 138, with the proportionalityconstant being determined by the area ratios of the current sourcetransistor with respect to the MOSFET diode transistor 133. As the termis used herein, “area ratio” refers to the well-known transistorwidth/length (W/L) ratio.

Operationally, differential output currents are developed by thedifferential pairs in response to control inputs a, b, c and d, eachdriving the gate terminal of a respective n-channel transistor 134, 135,136, and 137 configured as switches. N-channel switch transistor controlthe output current operation of the driver cell and determine the quantaof current defining the differential outputs.

For example, for matched current sources 131 and 132, each conducting acharacteristic current I, when control signals a and c are in a state soas to turn on corresponding switch transistors 134 and 136, whilecontrol signals b and d are in a state so as to maintain switchtransistors 135 and 137 in an off condition, the Ip output mode willdefine a current equal to 2×I, while In is equal to 0. Othercombinations will immediately suggest themselves to one having skill inthe art and can be easily determinable by merely turning the variousswitch transistors on or off along a programmed sequence until allpossible binary combinations of control signals states have beenexhausted. Thus, transistors 134, 135, 136, and 137, configured asswitches, control the output current operation of the line driver cellgenerated by the current sources.

As noted above, each individual current driver cell can be controlledfor either Class-A, Class-B or a combination of Class-A and Class-Boperation by operation of the Class-A and Class-B driver control logiccircuitry 123 and 124 of FIG. 12B. With reference to the current drivercell 126 of FIG. 13, Class-A and Class-B operation of the driver cellwill now be described in connection with the following Table 1 and Table2.

In particular, Class-A operation of the line driver current cell ischaracterized by a constant common output current, without regard to theactual value of the differential output current of the cell.

TABLE 1 OUTPUT SIGNALS INPUT SIGNALS Diff. Com. a b c d Ip In Mode Mode1 0 0 1 1.0 * I 1.0 * I 0 2.0 * I 1 0 1 1 1.5 * I 0.5 * I 1.0 * I 2.0 *I 1 0 1 0 2.0 * I 0 2.0 * I 2.0 * I 1 1 0 1 0.5 * I 1.5 * I −1.0 * I 2.0 * I 0 1 0 1 0 2.0 * I −2.0 * I  2.0 * I

As illustrated in Table 1, given the particular binary states of thecontrol signals a, b, c and d, the common output current is seen to havea constant value equal to 2.0*I. For example, when control signals a andd are high while control signals b and c are low, the correspondingswitch transistors 134 and 137 are both in the on state, causing themeach to conduct the full value I of the current generated by therespective current sources 131 and 132. Accordingly, the outputs Ip andIn each take on a value of 1.0*I.

As illustrated in the second row of Table 1, when control signal c istaken high, thus turning on the second switch transistor 136 of thecorresponding differential pair, each of the transistors of the pairconduct one-half of the current I defined by the respective currentsource transistor (in this case, transistor 132). Thus, In exhibits avalue of 0.5*I, while the additional 0.5*I conducted by its mate in thepair is a reflected in the value of Ip. Thus, Ip exhibits a value of1.5*I. The remaining combinations of binary states of the controlsignals a, b, c and d necessary to maintain a common output currentvalue of 2.0*I will be evident to those having skill in the art uponexamination of the remaining entries with Table 1. Since the outputcurrents (Ip and In) may take on only five values (0, 0.5*I, 1.0*I,1.5*I and 2.0*I), all that remains is to ensure that the absolute valuesum of the two currents is equal to, in this case, 2.0*I. As illustratedin Table 1, the algebraic sums of the currents define five particularvalues of differential output current, i.e., −2.0*I, −1.0*I, 0, 1.0*Iand 2.0*I as is expected.

Accordingly, a Class-A operated driver cell will be expected to have lowEMI emissions but consume a relatively higher amount of power due to theconstant common mode output signal. In Class-B operation, however, thedriver cell can be operated to produce the same degree of varyingdifferential current output signals but with a varying common-modecurrent output. In Class-B operation, power consumption is significantlyreduced at the expense of higher radiative emissions due to the varyingcommon-mode output current as illustrated in the following Table 2.

TABLE 2 OUTPUT SIGNALS INPUT SIGNALS Diff. Com. a b c d Ip In Mode Mode0 0 0 0 0 0 0 0 1 0 0 0 1.0 * I 0 1.0 * I 1.0 * I 1 0 1 0 2.0 * I 02.0 * I 2.0 * I 0 0 0 1 0 1.0 * I −1.0 * I  1.0 * I 0 1 0 1 0 2.0 * I−2.0 * I  2.0 * I

In one particular embodiment, such as might be implemented in atransceiver as depicted in FIG. 2, Class-A and Class-B logic circuits(123 and 124 of FIG. 12B) might be implemented to output control signalsa, b, c and d which define a truncated set of the differential andcommon-mode output currents illustrated in Tables 1 and 2, above. Asillustrated in FIG. 12B, the DAC control word outputs a pair of controlsignals In0 and In1 for each logic circuit and line driver cellcombination. Necessarily, each control pair of the DAC word is able totake on only four binary values (0:0, 0:1, 1:0 and 1:1).

FIG. 14A is a simplified schematic diagram of one particularimplementation of a Class-A logic circuit connected to receive an inputcontrol pair from the DAC word and generate the four driver controlsignals. FIG. 14B illustrates the corresponding logic table for derivinga, b, c and d control signals In0 and In1 in Class-A operation. TheClass-A logic circuit, indicated generally at 123, is characterized bymirror image circuits, each including a cross-coupled pair of two-inputNOR gates. The output of each NOR gate is buffered by an invertercircuit as are the DAC word control pair inputs. As illustrated in FIG.14A, each of the two input NOR gates has its cross-coupled inputconnected through a delay element ΔT which functions to prevent theoutputs of each mirror-image circuit from being at a logic low at thesame time.

As illustrated in the logic table of FIG. 14B, the DAC control pair In0and In1 takes on three binary values, i.e., 1:1, 0:1 and 1:0. For thefirst input value (1:1), only one switch transistor of each differentialpair of the driver cell of FIG. 13 is in operation. Thus, both Ip and Inare at a value of 1.0*I, the differential mode current is 0 and thecommon-mode current is 2.0*I. In the next input binary state, i.e., 0:1,a and c activate their respective switch transistors causing the Ipoutput to equal 2.0*I. Since b and d are low, their respective switchtransistors are off and In conducts no current. Thus, the differentialoutput current is 2.0*I and the common-mode output current is again2.0*I. Conversely, when the binary value of the DAC control pair isflipped from the previous state, i.e., 1:0, it will be understood that band d cause their respective switch transistors 135 and 137 to conductwhile the previous conduction pair 134 and 136 are off. Thus, Inconducts 2.0*I while Ip conducts 0 current. The differential current isthus −2.0*I while the common-mode current is again 2.0*I.

FIG. 15A is a simplified schematic diagram of a logic circuit adapted totake a DAC control word pair and develop the four control signals a, b,c and d in a manner suitable for operating the driver cell of FIG. 13 inClass-B mode. FIG. 15B is the corresponding logic table for deriving a,b, c and d control signals from In0 and In1 in a Class-B operationalmode. As depicted in FIG. 15A, In0 and In1 are buffered through invertercircuits to generate a, c and b, d, respectively.

The corresponding Class-B logic table in FIG. 15B illustrates thelogical states of the four driver control signals, the respective Ip andIn output drive by the driver cell in response to the control signals,the differential output current and common-mode output current withrespect to the same binary values of the DAC control pair (1:1, 0:1 and1:0) as was the case with FIG. 14B above. From the three inputconditions, it will be seen that only the first, i.e., 1:1, gives adifferent result from the Class-A case described above. The remainingtwo input conditions, i.e., 0:1 and 1:0, result in the same differentialmode and common-mode output current. In the first case, however, all ofthe four driver cell control signals are 0, thereby defining adifferential output current of 0 but with a corresponding common-modecurrent of 0 as well.

In accordance with the present invention, current driver cell controlsignals can be adaptively determined by Class-A and Class-B logiccircuits in order to choose a driver cell's operational mode in order tomeet conflicting requirements of power efficiency and reduced EMIemissions. In order to achieve the highest value of power efficiency,i.e., lowest power consumption, all of the current driver cells would beexpected to be placed in Class-B operational mode. Conversely, for thelowest EMI emissions configuration, it would be expected that all of thecurrent driver cells would be configured to operate in Class-A mode. Intypical application conditions, a transceiver's transmit DAC would beexpected to have its current driver cells operating in a mixed Class-A/Bmode. For example, in nominal 10BASE-T operation, approximately 40percent of the cells (ten cells) would be configured to operate inClass-B mode, while 60 percent of the cells (fifteen cells) would beconfigured to operate in Class-A mode. If the transceiver wereanticipated to operate according to the Tx standard, i.e., 1.0 voltswings, ten of the cells would be typically configured to operate inClass-A mode while the remaining fifteen cells would be disabled.

Disabling a particular cell would only require that that cell be placedin Class-B operational mode and the DAC control word pair (In0 and In1)would be set at a binary value so as to put all of the driver cellcontrol signals a, b, c and d in a low state. In the exemplaryembodiment, In0 and In1 would be asserted as 1:1. Once all of thecurrent cell control signals are in a low state, the correspondingcurrent cell conducts no current, effectively disabling that cell.

It should be noted that the current driver cells are topologicallyidentical, thus the same current cell is used whether the system is inClass-A or Class-B operational modes. There is therefore noincompatibility between Class-A and Class-B outputs. Further, it shouldbe understood that any number of current driver cells can be configuredto operate in Class-A or Class-B modes by merely programming a controlPLA to issue the appropriate select signals to the transmitter. Thedriver cells are therefore fully adjustable and the mix of Class-A andClass-B modes will depend solely on the application desired for thetransceiver. For example, notebook computer applications have a greatdeal of sensitivity toward power consumption while relegating EMIemissions to a secondary consideration. Since notebook computers arebattery operated and have a limited power supply lifetime, a transceiveroperating in such an environment would be configured to operateprimarily in Class-B mode.

Conversely, in an enterprise application, such as a wiring closet, thetransceiver would be configured to operate primarily in Class-A mode inorder to reduce EMI emissions. Power consumption considerations aretypically secondary in such applications.

A transmitter constructed according to the adaptively configurableClass-A/Class-B circuitry is further advantageous in that the same DACcontrol word (In0 and In1) is used to define the differential signaloutput in both the Class-A and the Class-B modes, as illustrated inFIGS. 14B and 15B. Since the same current cell is used in both cases,and since the DAC control word remains the same, the system isinherently seamless as a cross-mode platform. No complex decision logic,or multiple DAC decoder architectures are required.

To reduce the undesirable harmonics of the output signal, an analogdiscrete-time filter 9 is integrated with the DAC line driver 36 inaddition to the interpolating digital filter 33 as shown in FIG. 3.Referring now to FIG. 16, each DAC line driver cell 126 is capable ofproducing ½ the differential output current signal as well as the fulldifferential output current signal. The full differential output currentis generated by certain combinations of the class-A/class-B controlsignals a, b, c, and d as shown in rows 3, and 5 of table 1 and rows 3,and 5 of table 2. The half differential output current is generated bycertain combinations of the class-A/class-B control signals a, b, c, andd as shown in rows 2 and 4 of table 1 and rows 2, and 4 of table 2. Thecontrol signals a, b, c, and d are derived from the ROM 31 outputsignals.

For each output sample, the line driver control logic 162 drives thedriver cells such that for the first segment of the drive period 166 ofT/N, the cell produces ½ the differential output current signal 165. Forthe second segment of the drive period of T/N, the cell is driven by theline driver control logic 162 to produce the full differential outputcurrent signal 164. In one embodiment of the present invention, thedelay cell 161 generates the two segments of the drive period.

FIG. 17 shows one implementation of the delay cell 161. An inverter isformed at the input stage by MP1 and MN1 MOSFETs. The current throughthis inverter is limited by MOSFETs MP0 and MN0 biased by BIASP andBIASN, respectively. This limited supply current slows down theinverter. A capacitance is formed by the two MOSFETS MP2 and MN2 tofurther delay the output of the input stage inverter. The delayed outputof the input inverter, is then inverted by MN3 and MP3 MOSFETS to formthe OUT signal.

The line driver control logic 162 utilizes an accurate time referencesuch as a time-accurate delay circuitry 161 or a PLL, such as the oneshown in FIG.11, to drive the line driver cell 126 to either its fullamplitude or half of its full amplitude. The currents for each linedriver cell 126 are added at node 163 to generate the output signal ofthe transmitter. In a preferred embodiment, the first time segment andthe second time segment are equal to T/2. As a result, the analogdiscrete-time filter applies nulls to the output spectrum at oddmultiples of the interpolation rate, i.e., N/T, 3*N/T, 5*N/T . . . . Thefirst null reduces the image energy around N/T thus providingsignificant reduction in EMI emissions. For a 20 MHZ digital data inputrate and an interpolation rate of eight, the first harmonic at the DACoutput is at 160 MHZ. This can be represented by a sinusoid:A=Sin(2Π.160 MHZ. t). After the discrete time filtering at every T/2(i.e., 3.125 ns), the first harmonic is represented by a summation oftwo sinusoidal signals: A′=½ Sin(2Π.160 MHZ. t)+½ Sin(2Π.160 MHZ.(t+3.125 ns)). After expanding this equation, all the terms cancel outeach other, resulting in a null signal. However, for even multiples of160 MHZ (N/T) (e.g., 320 MHZ), the terms do no cancel out each other.

FIG. 18 depicts a magnified view of signal 181 (the dotted lines) andsignal 182 (solid lines) that the result of performing the analogdiscrete time filtering on the signal 181. As displayed by signal 182 inFIG. 18, the effective result achieved by discrete-time filtering ofsignal 181 is similar to interpolation or over-sampling by 2 by adigital filter. However, this technique is performed with less circuitcomplexity which results in reduced silicon area and lower cost.

FIG. 21A shows an example of a 10Base-T sinusoidal input signal runningat 10 MHZ. The resulting discrete-time filtered signal is shown in FIG.21B that has smoother edges resulting in a reduction of EMI emission.

As illustrated in FIG. 19, in one embodiment, a pair of capacitors, C1and C2, are added to the outputs of the line driver 36 in 10Base-T modeto provide additional high frequency filtering. The capacitors can beeither external (discrete) capacitors or on-chip capacitors as shown inFIG. 20. Each integrated capacitors of FIG. 20 is formed by connectingthe sources and drains of the respective MOSFET 191 or 192 together toform the bottom plate of each respective capacitor. A resistor (192 or194) is connected in parallel across each formed capacitor as shown inFIG. 20. The top plate of each capacitor in FIG. 19 and FIG. 20 isconnected to one of the two differential DAC outputs, respectively.

A MOSFET switch (193 or 196) is connected to the bottom plate of eachcapacitor and ground (VSS). A control signal, 10Base-T mode, controlsswitch 193 and switch 196. In 10Base-T mode, the switches are turned onconnecting the bottom plate of each capacitor to ground (VSS), thusactivating the capacitors. This creates a first-order filter at the DACoutput comprising the capacitor and the resistive component of thetransmission load. The first-order filter provides high frequencyfiltering for the differential output signal as well as any common-modesignal generated by the DAC.

In 100Base-TX or 1000Base-T where tighter output return loss is needed,the switches are turned off. The bottom plate of each capacitor is leftfloating, having a high impedance connection to ground (VSS) through theoff-impedance of the switch. This mode disables the first-order filterand preserves the wide-band high output impedance of the DAC.

The transmit signal cancellation circuit 5 of FIG. 1 incorporates firstand second replica transmitters, each of which are connected to andoperatively responsive to a digital word representing an analog signalto be transmitted. The first replica transmitter is coupled to thereceive signal path and develops a voltage mode signal which is equal tobut opposite in phase of a voltage mode portion of the transmit signal.The second replica transmitter is also coupled to the receive signalpath and develops a current mode signal having a direct phaserelationship with the transmit signal. The voltage mode and current modesignals are combined with the transmit signal on the receive signal pathand, in combination, cancel voltage and current mode components of thetransmit signal that might appear at the inputs of the receiver duringsimultaneous transmission and reception. In one particular aspect of theinvention, the main transmitter and the first and second replicatransmitters are constructed as current mode digital-to-analogconverters.

FIG. 22 depicts a semi-schematic, simplified block diagram of onearrangement of an integrated transceiver, including transmission signalcancellation circuitry in accordance with the present invention. Theintegrated transceiver is so termed because it is implemented as asingle integrated circuit chip. However, the transceiver is conceptuallyand functionally subdivided into a transmitter section 220 a and areceiver section 220 b connected to communicate analog bidirectionaldata in full duplex mode over unshielded twisted pair (UTP) wiring, suchas might be encountered in a typical local area network (LAN)architecture. In the exemplary embodiment of FIG. 22, the transmittersection 220 a and receiver section 220 b are coupled to a UTPtransmission channel through a line interface circuit 214 which providesDC offset cancellation, and the like between the transceiver signal I/Oand a twisted pair transmission channel 4.

In accordance with practice of principles of the invention, thetransceiver's transmit section 220 a is implemented to include a maintransmit digital-to-analog converter (TX DAC) 227 connected to receive adigital transmit signal and convert that signal into positive andnegative analog current mode signals suitable for transmission over thetwisted pair transmission channel 4.

In like fashion, the receiver section 220 b receives positive andnegative analog current mode signals from the transmission channel andconverts them into a digital representation in a receiveanalog-to-digital converter (RX ADC) circuit 215. Followinganalog-to-digital conversion, receive signals are directed to downstreamcircuitry in which digital representation of the receive signal isdemodulated, filtered and equalized by digital signal processing (DSP)circuitry as described in connection with FIG. 2. Prior to digitalconversion, the analog receive signal may be pre-processed by analogfront end circuitry 57 which is often adapted-to condition and analogreceive signal to a form suitable for conversion by the receive ADC 215.

Front end circuitry 57 might suitably include a high pass or a band passfilter configured to remove a certain amount of noise and interferencefrom a raw analog receive signal. Band pass filtration is oftenimplemented in architectures where the transmission channel issubdivided into a number of different pass bands each adapted to carrycertain types of intelligence. Band pass filtration thus allows onlysignals occurring in desirable portions of the channel spectrum to bedirected to the receive ADC 215 for conversion and further signalprocessing.

Analog front end circuitry 57 might also include automatic gain controlcircuitry, input buffer amplifiers, and the like, with variouscombinations being implemented depending on how the particular channelis configured and also depending on the input requirements of thereceive ADC 215, as is well understood by those having skill in the art.

From FIG. 22, it is evident that the signal lines carrying the positiveand negative analog receive signals are coupled between the receiversection 220 b and the line interface circuit 214 in parallel with thesignal lines carrying the positive and negative analog transmit signals.Necessarily, analog signals being transmitted to a remote transceiversimultaneously with another remote transceiver's communicating an analogreceive signal to the receiver section 220 b, will be asserted both onthe transmit signal lines as well as on the parallel-connected receivesignal lines.

Accordingly, in the absence of any conditioning or cancellationcircuitry, an analog transmit signal will superpose over an analogreceive signal at the analog front end 57 and/or the RX ADC 215. Giventhe substantially greater signal to noise ratio (SNR) of a non-channelimpaired transmit signal to a receive signal which is subject to channelimpairment, leakage, echos, and the like, it is evident that such ananalog transmit signal would substantially perturb a receive signal,making analog-to-digital conversion and downstream signal processingsubstantially more difficult.

Signal conditioning or cancellation of the analog transmit signal fromthe analog receive signal path is accomplished by cancellation circuitrywhich is coupled into the transmit and receive signal paths at a 3-waysignal nexis between the transmit DAC 227, the receive ADC 215 and theline interface circuit 214. Cancellation circuitry suitably includes twoquasi-parasitic current mode digital-to-analog converters, termed hereina positive replica DAC 226 and a negative replica DAC 225, incombination with first and second cancellation resistors 228 and 229.The positive and negative replica DACS 226 and 225, respectively, are sotermed because of the relationship of their signal sense configurationswith respect to the positive and negative output signal lines of the TXDAC 227.

In the case of the positive replica DAC 226, its positive signal line iscoupled to the positive signal line output from the transmit DAC 227while its negative signal line is, likewise coupled to the negativesignal line of the transmit DAC. In the case of the negative replica DAC225, its positive signal line is coupled through cancellation resistor229 to the negative signal line output from the transmit DAC 227. Thenegative replica DAC's negative signal line is coupled throughcancellation resistor 228 to the positive signal line of the transmitDAC. Each of the DACs 227, 226 and 225 are coupled to receive the samedigital transmit signal, i.e., the signal intended for conversion by thetransmit DAC 227 and transmission over the channel 4 through the lineinterface circuit 214. Thus, the input to all of the DACs is anidentical signal.

In operation, the negative replica DAC 225 may be implemented as acurrent mode DAC and functions, in combination with cancellationresistors 228 and 229, to define a cancellation voltage, with equalvalue but opposite phase to the output defined by the transmit DAC 227.Because a negative replica DAC is likewise coupled, in reverse fashion,to the receive ADC 215, the cancellation voltage may also be thought ofas applied to the analog front end. Thus, voltage components of atransmit signal are removed from the receive signal lines prior to theirintroduction to the analog front end.

Because the cancellation voltage is developed by sourcing/sinkingcurrent through cancellation resistors 228 and 229, the excess currentssourced/sunk by the negative replica DAC 225 must also be compensated atthe output signal lines in order to ensure a proper output voltage atthe line interface circuit 214. The positive replica DAC 226 providesthe necessary current cancellation function by sinking/sourcing amatched, but opposite phase, current to that developed by the negativereplica DAC, thus resulting in zero excess current at the load,indicated in the line interface circuit 214 of FIG. 22 asseries-connected resistors 211 and 212, disposed between the positiveand negative output signal paths and including a common center tap to aground potential. It should be mentioned that the configuration of theline interface circuit illustrated in FIG. 22 is an AC equivalentcircuit. It will be understood that the circuit is able to berepresented in several DC configurations, which will exhibit the same ora substantially similar AC characteristic. Thus the line interfacecircuit 214 is exemplary.

In operation, cancellation resistors 228 and 229 define cancellationvoltages between the outputs of the transmit DAC 227 and the inputs tothe receive ADC 215 as a function of a bias current, developed by anadjustable bias circuit 224. The adjustable bias circuit 224 isconnected to the positive replica DAC and the negative replica DAC andprovides an adjustable bias current to each of the circuit components.The cancellation voltage developed by the cancellation resistors 228 and229 must cancel the output voltage of the transmit DAC 227 such that thesignal at the receive ADC terminals closely track only a signal receivedfrom a remote transmitter at the other end of the transmission channel4. The cancellation voltage across each cancellation resistor isnecessarily equal to the value of the cancellation resistor times thecurrent through that resistor (current sourced/sunk by the negativereplica DAC). In order to provide effective cancellation, thiscancellation voltage must be equal to the output voltage of the transmitDAC which is, in turn, equal to the current produced by the transmit DACtimes the load resistance at each terminal (resistor 211 or resistor 212in parallel with one half the distributed resistance value of thetwisted pair wire of the transmission channel).

In accordance with the exemplary embodiment, transmit DAC 227 isimplemented as a current mode DAC and defines an output current which isa function of a bias current, in turn defined by a bias circuit 221, thecurrent gain of the bias circuit 221 and the current gain of thetransmit DAC 227. Likewise, the cancellation voltage developed by thenegative replica DAC 225 is a function of the values of cancellationresistors 228 and 229, the current gain of the adjustable bias circuit224 and the current gain of the negative replica DAC 225.

FIG. 23 is a simplified circuit schematic diagram of the bias circuit221 of the transmit DAC 227. In simple terms, the bias circuit 221 mightbe described as a voltage follower in combination with a bias resistorwhich develops a stable reference current through one leg of a currentmirror. The stable reference current is mirrored to an output currenthaving a particular value defined by the stable reference current andthe transistor geometries of the devices defining the current mirror.

In particular, a reference voltage (VREF) is applied to the positiveterminal of an operational amplifier 231 whose output controls the gateterminal of an N-channel transistor 235. The N-channel transistor 235 isconfigured as a voltage follower, by having its source terminal fed backto the negative input of the operational amplifier 231. A current sourcetransistor 232 is coupled between the voltage follower device 235 and apower supply potential such as VDD so as to supply a source of currentto the voltage follower device 235. As will be understood by thosehaving skill in the art, the voltage follower device, in combinationwith the operational amplifier 231 function to impress a stable voltageat the device's source node which is equal to the value of the referencevoltage VREF applied to the positive terminal of the operationalamplifier 231. A bias resistor 222 is coupled between the voltagefollower's source node and ground potential, so as to define aparticular current value therethrough equal to the reference voltageVREF divided by the value of the bias resistor 222. This current ismirrored to a mirror transistor 233 which is configured with its gateterminal in common to the current source transistor 232. Thus, themirror transistor 233 conducts a proportional amount of current to thecurrent source transistor 232, with the proportionality governed solelyby the ratio of the sizes of the mirror transistor to the current sourcetransistor.

If, for example, with a given reference VREF the value of bias resistor222 were selected in such a way as to define a current of 1 mA throughcurrent source transistor 232, and if mirror transistor 233 wereconstructed to have a width over length (W/L) ratio of twice that of thesource transistor, mirror transistor 233 would define a bias current of2 mA at the bias circuit output 234. Thus, the bias current developed bybias circuit 221 will be understood to be a stable current which is afunction of VREF, the bias resistor 22 and the ratio of transistor sizesof the current mirror. The ratio of transistor sizes of the currentmirror determines the current gain of the mirror and is easilycalculable and adjustable during circuit design.

Turning now to FIG. 24, there is depicted a simplified transistorschematic diagram for the adjustable current bias circuit 224 of FIG.22. The construction and operation of the adjustable current biascircuit 224 is similar to construction and operation of the bias circuit221 described in connection with FIG. 23 above. An operational amplifier241 is operatively responsive to a reference voltage VREF and controlsthe gate terminal of an N-channel transistor configured as a voltagefollower 242 to mirror the reference voltage value at its sourceterminal. A bias resistor 223 is coupled between the source terminal andground potential in order to develop a reference current therethrough ina manner similar to the bias resistor 222 of FIG. 23. A current sourcetransistor 243 is coupled between VDD and the source terminal of thevoltage follower transistor 242 and mirrors the reference current toparallel-coupled mirror transistors 244 and 245. Mirror transistors 244and 245 each define a bias current at respective output nodes 247 and246 of the adjustable bias circuit 224.

In contrast to the bias circuit 221 of FIG. 23 above, the mirrortransistors 244 and 245 are each constructed to be ⅕ the size (have ⅕the W/L ratio) of the current source transistor 243. If the referencecurrent developed across bias resistor 223 was designed to have a valueof 1 mA, the current conducted by mirror transistors 244 and 245 wouldnecessarily have a value equal to about 0.2 mA. Thus, the current gainof adjustable bias circuit 98 would be in the range of about 0.2, whilethe current gain of 224 e bias circuit 221 would be in the range ofabout 2.0.

In a particular embodiment of the present invention, the bias currentsdeveloped by mirror transistors 244 and 245 are able to be adjusted tocompensate for variations in transmission line load in order to producea null transmission signal voltage at the inputs to the receive ADC.Bias current adjustment may be made by adaptively changing the value ofbias resistor 223 in order to adaptively modify the value of thereference current developed therethrough. Adjusting the value of thebias resistor 223 can be carried out internally by trimming the resistorat the time the apparatus is packaged as an integrated circuit, or byadaptively writing a control word to a control register that controlsthe configuration of a resistor ladder. Likewise, it will be understoodthat-adjustment may be made externally by coupling a potentiometer orvariable resistor in parallel with bias resistor 223.

Alternatively, bias current adjustment may be made by dynamicallychanging, or adjusting, the sizes of the mirror transistors 244 and 245as well as the size of the source transistor. In the present exemplarycase, where a 1:5 ratio between currents is desired, the current sourcetransistor might be constructed as an array of fifty (50) transistors,and each of the mirror transistors might be constructed as an array often (10) transistors. As changes in the current ratio become desirable,fuse-links coupling the transistors into the array might be “opened” byapplication of a current, thereby removing a selected transistor ortransistors from the array.

Adjusting a bias current by adaptively “trimming” transistors gives ahigh degree of flexibility and control to the actual value of thecurrent output by the circuit. Transistor trimming of transistorsconfigured in a series-parallel array allows incremental fine tuning ofcurrents, the precision of which is limited only by the number oftransistors in the array and the unit widths (W) and lengths (L) usedfor the elemental transistors.

Returning now to FIG. 22, it should be noted that the current gains ofthe transmit DAC 227, the positive replica DAC 226 and the negativereplica DAC 225 are all designed to be matched and identical. This isaccomplished by replicating the integrated circuit design of thetransmit DAC to the positive and negative replica DACS. Thus, since thetransistor layout and design parameters of all of the DACs are similarit would be expected that the performance characteristics, such as gain,of the DACs would be similar as well. In like fashion, the circuitdesign and layout of the bias circuit 221 is replicated in theadjustable bias circuit 224, with the exception of the transistorsizings of the mirror transistors. Thus, the current gain of theadjustable current bias circuit 224 is expected to proportionally trackthe current gain of current bias circuit 221 over the corners ofintegrated circuit manufacturing process variations. That is, if thegain of bias circuit 221 is skewed in one direction by a certainpercentage, the gain of the adjustable bias circuit 224 will be expectedto also vary in the same direction by approximately the same percentage.Accordingly, the ratio of the bias current developed by bias circuit 221to the bias currents developed by adjustable bias circuit 224 willremain substantially constant.

In accordance with the principles of the invention, the current gain ofthe adjustable bias circuit 224 is chosen to be substantially smallerthan the current gain of bias circuit 221, in order to minimize thecurrent and power requirements of the positive and negative replicaDAC's line driver circuitry. Accordingly, the values for thecancellation resistors 228 and 229 are selected so as to develop acancellation voltage equal to the transmit DAC output voltage, based onthe designed current gains. In other words, based on Ohm's law, thesmaller the output current, the larger the required cancellationresistors in order to produce a fixed cancellation voltage equal to thetransmit DAC output voltage.

Because the positive replica DAC 226 is closely matched in performancecharacteristics with a negative replica DAC 227, the current that thenegative replica DAC sources/sinks is canceled by a matched currentsunk/sourced by the positive replica DAC. This current cancellationresults in zero excess current at the transmit DAC output, leaving onlythe desired transmit signal at the line interface load.

In order to ensure stability of the voltage cancellation function overmanufacturing process parameter, power supply voltage and thermalvariations, the adjustable bias circuit resistor 223 and thecancellation resistors 228 and 229 are constructed from the samesemiconductor material (polysilicon, for example) and are laid out inproximity to one another so as to track each other over processparametric, power supply and/or thermal variations. In this manner,induced cancellation voltages across cancellation resistors 228 and 229,will be understood to be independent of process variations. Because thepositive replica DAC 226 is driven by the same adjustable bias circuit224 as the negative replica DAC 225, the cancellation currents developedby the positive replica DAC will be expected to closely track thecurrents developed through negative replica DAC 225.

One particular utility of the present invention may be found in itsability to produce a cancellation signal which is substantially a mirrorimage of a simultaneously asserted transmit signal and provide thecancellation signal at the input of a transceiver's receive ADC oranalog front end. The effectiveness of the present invention will bemore clearly understood with reference to the timing diagram of FIG. 25which illustrates the signal state at various nodes in the exemplarytransceiver circuit of FIG. 22. For example, the periodic signaldepicted at FIG. 25(a) might represent the source voltage developed by aremote transceiver at the other end of the transmission line which is tobe received by the local transceiver. The signal depicted at FIG. 25(c)might represent an analog transmit signal developed by the localtransmitter and which is simultaneously asserted to the line interfacecircuit and the transmission channel as the intended receive signaldepicted at FIG. 25(a). The signal illustrated in FIG. 25(b) representsthe signal that might be seen on the channel (4 of FIG. 22) and might bedescribed as a linear combination of the transmit signal (c) and thereceive signal (a) along with such impairments as are common in UTPtransmission channels.

The signal depicted at FIG. 25(d) represents the signal appearing at theinput to the analog front end or the receive ADC, after the transmitcancellation signal has been subtracted from the combination signal at(b). As can be seen from the waveform diagrams of FIG. 25, the receivesignal (d) has a substantially greater fidelity to the original signal(a) than the combination signal (b) appearing on the channel.

Notwithstanding its ability to effectively and accurately cancel localtransmit signals from a local receiver's input signal path, theinvention is additionally advantageous in that it obviates the need forcomplex and costly external magnetic hybrid circuits to interfacebetween a transceiver in a twisted pair transmission channel. Inparticular, as can be seen in FIG. 22, the line interface circuit 214,between the transceiver and the channel, can be simply implemented by apair of series coupled resistors and a relatively simple transformerelement (indicated at 213 in FIG. 22) which, in the present case, isneeded only to provide common-mode voltage rejection and DC isolationbetween the channel and the transceiver I/O.

Further, transmit signal cancellation circuitry and the line interfacecircuit are particularly suitable for implementation in a single chipintegrated circuit. The replica DACs and resistors are all constructedof common integrated circuit elements and can be implemented on a singlechip along with the remaining components of a high speed bidirectionalcommunication transceiver. In accordance with the invention, only thetransformer portion of a line interface circuit is contemplated as anoff-chip circuit element. Even though the exemplary embodimentcontemplates the transformer being provided off-chip, it will beunderstood by those familiar with integrated circuit design andfabrication that suitable transformers can be constructed fromintegrated circuit elements, such as combinations of spiral inductors,and the like, and still provide sufficient DC coupling between atransmission channel and an integrated circuit transceiver.

While the adaptive signal cancellation circuitry has been described interms of integrated circuit technology implementing a gigabit-typemulti-pair ethernet transceiver, it will be evident to one having skillin the art that the invention may be suitably implemented in othersemiconductor technologies, such as bipolar, bi-CMOS, and the like aswell as be portable to other forms of bidirectional communicationdevices that operate in full duplex mode. Moreover, the circuitryaccording to the invention may be constructed from discrete componentsas opposed to a monolithic circuit, so long as the individual componentsare matched as closely as possible to one another.

A multi-transmitter communication system may be configured fortransmitting analog signals over a multi-channel communication network.The system is constructed to incorporate M transmitters, each having anoutput for serving a transmit signal on a transmit signal pathelectrically coupled between each communication channel and the outputof the respective transmitter. A timing circuit is electrically coupledto each transmitter for providing the required timing signals for eachtransmitter. The timing signals for the transmitters define a clockdomain that is staggered in time resulting in a respective phase shiftof the output signals of each transmitter. In one embodiment of thepresent invention, the timing signals are staggered in time forpredetermined time intervals to reduce aggregate electromagneticemission caused by signal images centered around integer multiples offrequency Fi of the M transmitters. M timing references staggered intime by 1/(Fi*M) are generated by the timing circuit to drive the outputof each of the M transmitters respectively.

Referring now to FIG. 26, an emission reduction technique for fourtransmitters is shown. In one embodiment of the present invention, acommon time reference circuit 7 provides the required timing signals toall of the transmitters, however, the time reference to each transmitteris delayed by a predetermined period of time. The time referencestaggered delays, 116 a to 116 d, of each transmitter is chosen toreduce the aggregate EMI emissions of the system. This approach alsoreduces the noise from the system power supplies by requiring smallercurrent requirement at a given time. This technique can be extended tosystems with several transmitters such that the time reference to themultiple transmitters are staggered on a PCB or an IC chip using delaylines or delay logic. The time staggering signals can be derived, forexample, from a PLL as shown in FIG. 5.

Assuming an output sample frequency of Fi, images contributing to EMIemissions for each transmitter are centered around 1*Fi, 2*Fi, 3*Fi, . .. , the time references of M transmitters are staggered in time by1/(Fi*M) . This timing arrangement places nulls, in the aggregate EMIemissions, at 1*Fi, 2*Fi, 3*Fi, . . . except at frequency multiples ofM*Fi. This staggering technique reduces the EMI emissions caused byimages located around the null frequencies.

As an example, images of a single 10Base-T transmitter are located at160 MHz, 320 MHz, 480 MHz, . . . . For an application which implementsfour transmitters on a single chip, the time references are staggered by1.5625 ns (1/(Fi*M)). This reduces the aggregate EMI emissions of thesingle chip device at 160 MHz, 320 MHz, 480 MHz, 800 MHz, . . . but notat 640 MHz, 1280 MHz, . . . . FIG. 27 shows the image components of fourexemplary transmitters. The images are each shifted by 90 degrees inphase, and by 1.5625 ns in time. As illustrated by the timing diagram ofFIG. 6, the aggregate power of the images is zero.

For the above 10Base-T example, the aggregate image voltage of fourtransmitters, before any staggering, can be represented by:

V=Sin(2Π·160 MHZ. t)+Sin(2Π·160 MHZ. t)+Sin(2Π·160 MHZ. t)+Sin(2Π·160MHZ. t)−4 Sin(2Π·160 MHZ. t). However, after staggering the timingreference of each transmitter by 1.5625 ns (Δt), the aggregate imagevoltage is: V′=Sin(2Π·160 MHZ. t)+Sin(2Π·160 MHZ. (t+Δt))+Sin (2Π·160MHZ. (t+2Δt))+Sin(2Π·160 MHZ. (t+3Δt). The terms of this equation cancelout each other at 160 Mhz. The same cancellation effect occurs forimages at 320 MHz, 480 MHz, 800 MHz, . . . but not at 640 MHz, 1280 MHz,. . . . This technique can be implemented in any electronic subsystemincluding PCBs and IC chips.

The staggered timing signals can be accurately generated by a timingcircuit, such as a PLL that includes a Voltage Control Oscillator (VCO).FIG. 11 depicts a PLL used for generating the required staggered timingsignals for the multiple transmitter configuration in one embodiment ofthe present invention. Other techniques for generating timing referencesignals known in the art of circuit design may also be used to generatethe required staggered timing signals.

The present invention is additionally advantageous in that it can beconfigured to operate between and among various Ethernet transmissionstandards. In particular, by merely disabling or re-enabling groups ofmemory arrays and current driver cells, the transmitter according to theinvention can operate under 10BASE-T, 100BASE-T, 100BASE-Tx and1000BASE-T standards seamlessly. Thus, a single integrated circuittransceiver is able to perform a multiplicity of roles under a varietyof conditions in a seamless and flexible manner.

Neither are the principles of the invention limited to the particularEthernet standards discussed above. As standards evolve, differingdigital filtering and output voltage swing requirements are easilyaccommodated by the present invention by changing the contents of thememory device, and changing the “width” of the DAC control word and thenumber of driver cells to capture the new requirements. Nor is theinvention limited by the number of cells making up a voltage step. DACresolution and accuracy can be further enhanced by defining “quarter”cells, and the like, and making appropriate changes to the decoder andswitching logic sections.

It will be recognized by those skilled in the art that variousmodifications may be made to the illustrated and other embodiments ofthe invention described above, without departing from the broadinventive scope thereof. It will be understood therefore that theinvention is not limited to the particular embodiments or arrangementsdisclosed, but is rather intended to cover any changes, adaptations ormodifications which are within the scope and spirit of the invention asdefined by the appended claims.

1. A low aggregate electromagnetic emission transmission systemcomprising: a plurality of analog transmitters adapted to concurrentlydrive a plurality of twisted wire pair transmission channels; means forswitching each of the transmitters on, to generate an analog transmitsignal, at a different time from the other transmitters such that theconcurrent output signals of the plurality of transmitters arephase-shifted with respect to one another, the interval betweenswitching analog transmitters being inversely proportional to the numberof transmitters.
 2. The system of claim 1, wherein the interval betweenswitching transmitters is also inversely proportional to a transmissiondata rate.